E-beam high voltage switching power supply

ABSTRACT

A high-power power supply produces a controllable, constant high voltage  put under varying and arcing loads. The power supply includes a voltage regulator, an inductor, an inverter for producing a high frequency square wave current of alternating polarity, an improved inverter voltage clamping circuit, a step up transformer, an output rectifier for producing a dc voltage at the output of each module, and a current sensor for sensing output current. The power supply also provides dynamic response to varying loads by controlling the voltage regulator duty cycle and circuitry is provided for sensing incipient arc currents at the output of the power supply to simultaneously decouple the power supply circuitry from the arcing load. The power supply includes a plurality of discrete switching type dc--dc converter modules.

The United States Government has rights in this invention pursuant toContract No. W-7405-ENG-48 between the United States Department ofEnergy and the University of California for the operation of LawrenceLivermore National Laboratory.

This is a Division of U.S. application No. Ser. No. 07/867,639 filedApr. 13, 1992 now U.S. Pat. No. 5,418,707.

BACKGROUND OF THE INVENTION

1. Field Of The Invention

The field of the present invention generally relates to high power solidstate power supplies. More particularly, the field of the presentinvention relates to a high power solid state power supply for producinga controllable, constant high voltage output under varying and arcingloads suitable for powering an ion source, such as an electron beam gunin a vacuum furnace, or an electron beam gun used in the vaporizer of alaser isotope separation system, or a plasma sputtering device or thelike.

2. The Prior Art

An electron beam gun is used in a vacuum furnace system, or the like,for providing a high intensity beam of electrons to bombard a targetmaterial. The electron gun is typically disposed in an evacuated chambertogether with the target material. The electron gun or E-beam gunusually includes a source of electrons, such as a heated cathode orfilament, and a grounded accelerating anode. The cathode is maintainedat a high negative potential with respect to the anode to establish ahigh electrostatic field for accelerating the electrons. A magneticfield may typically be provided for directing the electrons onto thetarget material.

During bombardment of the target material by the electron beam, variousionized materials are emitted. The presence of such materials ofteneffects a substantial decrease in the voltage withstand capabilitybetween the various parts of the electron beam gun and other elements.This may result in arcing between the electron gun parts and otherstructures. Arcing causes a substantial increase in the electron guncurrent and may result in damage to the electron gun structure andsurrounding elements. Arcing may also cause damage to the powercircuitry driving the electron gun.

In high power and high performance applications, such as the vaporizerin a laser isotope separation system, physical spacing between theE-beam gun, surrounding components, and target materials is relativelysmall. As a result, the E-beam gun may arc to ground frequently. Toavoid damage and to achieve long lifetimes, it is essential that theenergy stored in the power supply output capacitance be small and thatthe so called power supply let through energy during arcing be small. Inaddition, the close physical spacing causes a greater chance for theelectron beam to impinge on adjacent components and structures duringsteady, non-arcing operation. To avoid this, it is important that thepower supply output voltage be accurately controllable with low ripplecontent.

Conventional thyristor controlled power supplies are inadequate for highpower and high performance electron beam gun applications. Thyristorcontrolled power supplies generally operate at 60 Hz line frequency andgenerate significant output voltage ripple or require substantial outputcapacitance to reduce the ripple to acceptable levels. If gun arcs arefrequent, the output capacitance may result in excessive accumulatedenergy discharge into the gun or surrounding components and result in ashort lifetime. Thyristor controlled power supplies also have arelatively slow dynamic response which results in further energy letthrough to the gun during the arc and slow ramp up after the arc isextinguished. Thyristor controlled power supplies have a relatively poorinput power factor and generate high input harmonics. This causessubstantial cost increases in the 60 Hz utility power system in largepower applications. Thyristor controlled power supplies are alsophysically large because the transformer and filter components operateat 60 Hz and the lower harmonics of 60 Hz. This is an important factorin capital equipment costs where large numbers of power supplies areused.

Conventional power supplies utilizing series pass tetrode vacuum tubeseliminate many of the deficiencies of the thyristor controlled powersupply. The regulating characteristics of the tetrode vacuum tube can beused to produce very low output ripple voltages without requiringsignificant output capacitance. The regulating characteristics alsopermit a diode rectifier front end to be used which greatly raises theinput power factor and reduces the input line harmonics. The currentlimiting tube characteristics, the high speed control capability of thetetrode grid, and the low output capacitance provide excellent responseto gun arcs resulting in low energy into the gun and fast recovery afterthe arc extinguishes.

However, conventional tetrode vacuum tube E-beam power supplies haveserious deficiencies of their own. The efficiency of this type of powersupply is 80% or less compared to approximately 95% for thyristorcontrolled power supplies. This is because the tetrode must dropsubstantial voltage continuously for it to regulate properly. Tetrodevacuum tubes also wear out due to the filament breaking and to thechemical breakdown of the coating on the cathode which causes thecathode to lose its ability to emit electrons. As a result, the tetrodevacuum tube is a substantial maintenance expense item having to bereplaced at least every 10,000 hours.

Power supplies which use switch-mode dc--dc converters operating at 10kHz and above have the potential to eliminate the deficiencies of theconventional thyristor controlled and series-pass tetrode type powersupplies. Power supplies which employ switch-mode dc--dc converters arecompact because of smaller transformer and filter components, operatewith a diode rectifier input for high input power factor, are efficientbecause they do not operate as linear regulators, require lowmaintenance because they are all solid state, and can have good dynamicresponse because they operate at high frequency.

One type of switching dc--dc converter useful for high powerapplications above 10 KHz with arcing loads is the series resonant type.Power supplies which use series resonant type dc--dc converters have aninput rectifier and filter to produce a dc voltage, an inverterconsisting of thyristors and a resonant network to produce highfrequency current, transformer for producing the desired output voltagelevel, and a rectifier and filter to produce dc for application to theload. This is a well known type of power supply which has been appliedto E-beam guns (U.S. Pat. No. 3,544,913, issued Dec. 1, 1970). The majordeficiencies in this type of power supply for high performance E-beamapplications are the amount of energy stored in the output filtercapacitance and the inability to turn off power to the load until theresonant network reverses polarity. The output current of the inverteris sinusoidal and a substantial capacitance is required afterrectification to obtain satisfactory output voltage ripple even thoughthe inverter operates above 10 KHz. The dc--dc converter also continuesto provide current to the load after an arc occurs until the resonantnetwork commutates the thyristors. Although superior to the conventional60 Hz thyristor type power supply with respect to energy dissipated intothe gun during arcing, it is inferior to the series pass tetrode typepower supply, and is not adequate for high performance power suppliesfor E-beam guns.

Another type of switching dc--dc converter useful for low powerapplications up to a few kilowatts and arcing loads is the currentsource, pulse-width-modulated type. This dc--dc converter consists of avoltage regulator, inductor, non-regulating inverter, transformer,output rectifier, and output filter capacitor as described in U.S. Pat.No. 3,737,755, issued Jun. 5, 1973. The inductor and inverter describedin this referenced patent produce a square current waveform to theoutput rectifier and filter which allows a small output filtercapacitance to be used and therefore low energy to the load during loadarcs. However, the inverter voltage clamping means is inadequate forhigh power applications. This is because the inverter is relativelydistant from the input filter capacitor in high power applications whichresults in substantial inductance in the clamping network and excessivevoltage spikes across the inverter transistors.

As described above, problems exist with conventional power supplies andwith switching power supplies for high power and high performance ionsources and specifically, electron beam guns. In summary, conventionalthyristor power supplies have high output capacitance, slow dynamicresponse to arcing, and poor input power factor. Series pass tetroderegulator type power supplies have relatively low efficiency andsubstantial maintenance expense related to the vacuum tube. Both ofthese conventional types of power supplies are also physically large.Switching power supplies using series resonant type dc--dc converterssolve many of the problems associated with conventional power suppliesbut still have excessive output capacitance and too slow a response toarcing. Switching power supplies using current source,pulse-width-modulated type dc--dc converters as described in the priorart potentially meet the E-beam gun power supply requirements but do notoperate at high power levels.

Accordingly, it is an object of the present invention to provide animproved current source, pulse-width-modulated type dc--dc convertersuitable for operation at 100 kW or more.

It is another object of the present invention to provide a power supplywhich is modular with one or more dc--dc converter modules of identicaldesign rated at 100 kW or more used to achieve the required outputpower.

It is also an object of the present invention to provide a power supplywhich has tight voltage regulation and low output voltage ripple forprecise beam control as well as small output capacitance for smallenergy into the load during load arcs.

A further object of the present invention is to provide a power supplywhich is current limited during an arc, which cuts back power to zerowithin a few micro-seconds or less after an arc is initiated, and whichthen ramps power back on in several milli-seconds after the cutbackinterval.

A still further object of the present invention is to provide a powersupply which operates without excessive voltage transients or cablereflections with cable lengths between the power supply and the load of100 feet or more during and after load arcing.

SUMMARY OF THE INVENTION

In accordance with the present invention, a solid state switching powersupply is provided for converting a utility supplied ac input voltageinto a high voltage dc output suitable for driving an electron gun for ahigh power and high performance application such as in an atomic vaporlaser isotope separation process or the like. The present invention alsomay be useful for applications such as radar systems, which may behaveas an arcing load.

The present power supply comprises an input diode rectifier and filterfor converting the ac input voltage to an unregulated dc voltage. Aplurality of switching type dc--dc converter modules, each rated at 100kW or more, are connected with their inputs in parallel and theiroutputs in series and used to convert this unregulated dc voltage into aregulated high dc voltage output.

The dc--dc converter modules are an improved version of the well knowncurrent source, pulse-width-modulated type. Each converter consists ofan input decoupling network, an input capacitor, a voltage regulator, asquare wave inverter, a step up transformer, and an output rectifier andfilter. The input decoupling network functions together with the inputcapacitor to eliminate interaction between modules and to preventsignificant high frequency current in the cables from the inputrectifier and filter. This feature permits parallel operation of themodules from one dc source. Insulated gate bipolar transistors (IGBTs)are used as the switches in the voltage regulator and inverter becauseof their high power capability and fast switching speeds. An improvedinverter clamping network minimizes stray inductive loops and permitstight control of the inverter IGBT voltages during inverter outputpolarity transitions. Both of these features permit module operationwith inverter switching frequencies of 10 kHz and above and output powerlevels of 100 kW and above.

Each dc--dc converter module has a control scheme to permit operationwith widely varying and arcing loads. A feedback loop regulates themodule output voltage by controlling the on/off ratio of the voltageregulator switches. A feedback signal representing the output voltage isgenerated using an output simulator circuit fed from a one turn sensewinding on the step up transformer and a dc current sensor on the moduleoutput lead. This feature permits generation of a feedback signal atground level, isolated from the high voltage, which reproduces thedynamics of the module output power circuitry and the e-beam gun load.The control scheme also blocks power flow to the load during arcing.Output currents above normal operating levels are detected by acomparator circuit which triggers a cutback timer circuit. This cutbacktimer simultaneously turns off both regulator switches and turns on allfour inverter switches for a predetermined time interval. This featurepermits rapid extinguishing of the arcing in the load.

The overall power supply output is formed by connecting the outputs ofthe modules in series and then connecting to an output decouplingnetwork and a transmission line matching impedance before connecting tothe transmission line to the load. The output decoupling networkprevents the module output filter capacitors from shorting out the linematching impedance. This feature keeps overvoltage transients to aminimum at the power supply output and at the load during load arcing.

The overall power supply includes control circuitry for phasing thetiming of the switching of the voltage regulator and inverter in eachmodule with respect to adjacent modules by 360° /n for n total modules.This feature reduces the output capacitance needed to achieve a lowoutput voltage ripple and therefore reduces the energy delivered to theload during load arcing. Circuitry is also included for controlling thepower supply output voltage. An overall voltage feedback loop producesan error signal which serves as the voltage setpoint to the individualmodule feedback loops. A ramp generator controls the rate of rise of theoutput voltage during turn on and after cutbacks.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1A and FIG. 1B together are an overall circuit diagram of the powersupply in accordance with the present invention. It is shown with fourdc--dc converter modules.

FIG. 2 is a circuit diagram of a single module.

FIG. 3 is a partial circuit diagram of a single module with anequivalent representation of the transformer primary.

FIG. 4 is a timing diagram showing the operation and waveforms of theinverter clamping circuit.

DETAILED DESCRIPTION

As shown in FIG. 1A and FIG. 1B (hereinafter jointly referred to as FIG.1), an ac voltage, typically 480 volts, three phase, is applied to arectifier/filter circuit 100. Rectifier/filter circuit 100 provides ameans for converting the three phase ac input voltage to a filtered dcoutput voltage on positive node 101 and reference node 102,respectively. The rectifier/filter consists of six diodes 103 arrangedin a full wave bridge circuit. The output of the full wave bridgecircuit connects to inductor 104 to provide smoothing of the currentripple. The inductor 104 then connects to a series connected networkformed by damping resistor 107 and filter capacitor 108. For an outputrating of 400 kW, the inductor 104 is typically 500 microhenries (μH),the damping resistor 107 is typically 0.5 ohms, and the capacitor 108 istypically 8,000 microfarads (μF). The rectification of the ac voltageand the filtering of the voltage ripple are accomplished in a knownmanner.

In accordance with one aspect of the present power supply, a pluralityof modular dc--dc converter circuits or modules 112-1, 112-2, 112-3, . .. 112-n, are connected in parallel to the output leads, nodes 101 and102, respectively of the rectifier/filter 100. Each dc--dc convertermodule 112-1 through 112-n includes a voltage regulator circuit means114 for producing a source of pulsating voltage of controlled dutycycle; an inductor 128 for converting the pulsating voltage into adirect current, an inverter circuit means 118 for generating a highfrequency square wave of alternating polarity from said direct current,a transformer means 150 for isolation and step up of the inverteroutput, and an output rectifier/filter circuit means 160 for rectifyingand filtering the high frequency square wave current to produce a highdc output voltage.

Each voltage regulator circuit 114 is associated with a correspondingdecoupling network 116 and input capacitor 125. The decoupling networkcomprises a 0.25 ohm resistor 120 and a 40 microhenry (μH) inductor 122connected in parallel and having a connection with the positive inputlead 101 from the rectifier 100. The input capacitor 125 is typically200 microfarads (μF) and one side connects to the node formed by thedecoupling network and the positive inverter bus 141a. The other side ofinput capacitor 125 connects to the negative input lead 102 from therectifier/filter 100. The decoupling network, inductor 122 and resistor120, provides an impedance at high frequency which forces the currentsurges drawn by regulator 114 to pass through capacitor 125 and notthrough the rectifier/filter 100 or the parallel connected dc--dcconverter modules.

Each voltage regulator circuit 114 comprises two insulated gate bipolartransistors (IGBTs) 124 in parallel having their emitters connected withthe negative lead of input capacitor 125 and having their collectorsconnected with the anode of a free wheeling diode 126 and with a firstlead of a 500 microhenry (μH) inductor 128, respectively. The presentdevice is not limited to IGBTs. Any gate controlled switching means ofsuitable current and voltage ratings and switching speed may be used.For simplicity, such devices will be referred to as IGBTs. IGBTs 124a,124b are activated alternately by control signals applied to theirgates. Both IGBTs 124a and 124b are activated at 10 kHz but phaseshifted 180° with respect to each other. This results in a net 20 kHzswitching frequency for the pair. This technique eliminates currentsharing difficulties which might occur if both IGBTs 124a and 124b wereactivated simultaneously. The activation of IGBTs 124a, 124b enablecurrent to build up and to decay through inductor 128 in such a mannerthat the output voltage of the dc--dc converter module 112 iscontrolled. The activation of IGBTs 124a, 124b causes a rectangular dcvoltage waveform of from 0 to 650 volts to appear across diode 126. Thepulsating dc voltage from 0 to 650 volts across free wheeling diode 126is smoothed out to a low ripple direct current by inductor 128. Theabove values stated for the decoupling network 116 components, inputcapacitor 125, and inductor 128 are for a 100 kW module output ratingand for a switching frequency of 10 kHz for each voltage regulator IGBT124a and 124b.

The dc output voltage from the voltage regulator 114 is passed throughinductor 128 to produce a substantially smooth direct current which isthen applied to an inverter means 118 for converting the dc to a highfrequency square wave current of alternating polarity, that is, a highfrequency ac. Current is supplied from the node formed by the inputdecoupling network 116 and input capacitor 125 to the inverter positivebus 141a and returned from the inverter negative bus 141b throughinductor 128 to the voltage regulator 114. Inverter 118 comprises aplurality of insulated gate bipolar transistors (IGBTs) connected inseries and configured in two branches which are in turn connected inparallel, forming a bridge circuit. Here also, any gate controlledswitching means suitable for high current applications may besubstituted for IGBTs. Preferably, four IGBTs 140 are used, two seriesconnected IGBTs in each parallel branch, which are activated in diagonalpairs. There are two IGBTs for each ac line. It will be appreciated bythose skilled in the art that two ac lines are needed to drive anassociated single phase transformer.

The IGBTs 140 are activated in alternate, diagonal pairs by controlsignals applied to their enable leads in a manner well known to thoseskilled in the art, so as to produce a square wave of alternatingpolarity. The high frequency square wave current is applied through theprimary winding of an associated transformer 150. The IGBTs are able todevelop the needed high frequency power due to their high switchingspeeds, high current capability, and high breakdown voltage. The highfrequency current through the primary is inductively coupled to thesecondary of the transformer 150. A transformer 150 is part of eachcorresponding dc--dc converter module 112-1 through 124-n, and providesa means for stepping up the high frequency ac voltage applied to theprimary in order to produce the high voltage needed for powering theelectron gun. The transformer 150 also provides electrical isolation forthe output of each dc--dc converter module which permits the outputs tobe connected in series. The leads of the secondary of transformer 150are then applied to an output rectifier/filter circuit means 160 forconverting the high frequency alternating current to a smooth high dcvoltage. The output rectifier/filter circuit 160 consists of a singlephase, full wave bridge rectifier which connects to a parallel 0.05microfarad capacitor 161 and a parallel damping network formed by a 2400ohm resistor 163 in series with a 0.15 microfarad capacitor 162. Theabove values are for a four module power supply rated at 400 kW and 50kV output with a ±0.5% peak to peak output voltage ripple, and eachinverter operating at 10 kHz. The output rectifier/filter circuit means160 operates in a well known manner.

Referring to FIG. 1, an output decoupling network 170 comprises a 100ohm resistor 172 in parallel with a 500 microhenry (μH) inductor 174.One side of the output decoupling network 170 is connected to the highvoltage side of the series connected dc--dc converter module outputs.The other side of the decoupling network is connected to the node formedby the high voltage transmission line to the load, the line matchingnetwork consisting of a 50 ohm resistor 175 and 0.01 microfaradcapacitor 176 connected in series, and the high voltage side of thevoltage feedback divider 180. For the sub-microsecond times in which aload-arc is initiated and transmission line reflections occur, theimpedance of the decoupling network is much larger than the impedance ofthe line matching network. As a result, during arcing the transmissionline is matched by its characteristic impedance and is not shorted outby,the output filter capacitance of the dc--dc converter modules. Thisresults in small over-voltage transients at the load and power supply.The typical component values listed above are for an output of 50 kV at400 kW and with a transmission line length of 100 feet.

It will be apparent to those skilled in the art that the outputdecoupling network 170 can be divided evenly and distributed into eachdc--dc converter module. The function of the decoupling network will notbe changed. In some applications this is the preferred embodiment.

It will also be appreciated by those skilled in the art that the outputsof the dc--dc converters can be connected in parallel instead of inseries. This requires the output decoupling network 170 to be placed inthe output of each converter module to avoid interactions between theoutputs of each module.

In accordance with the present device, the inverter IGBTs 140a, b, c, dhave their on times synchronized with the on times of the IGBTs orswitching means 124a, b of each voltage regulator circuit 114 in eachdc--dc converter module 112. The transition of inverter output currentpolarity occurs when one diagonal IGBT pair in the inverter bridge turnsoff and the opposite diagonal IGBT pair conducts. This time occurs atthe end of conduction of either IGBT 124a or 124b in the voltageregulator circuit. This synchronization permits the phasing of themodules described below to be realized.

It will be appreciated that the dc--dc converter modules, shown assections 112-1, 112-2, 112-3, . . . 112-n in FIG. 1, comprise identicalmodular power supply circuits which are interchangeable. The inputs toeach dc--dc converter module 112 are linked in parallel and the outputsare linked in series with an adjacent dc--dc converter module 112.

The IGBTs 124a, b of each voltage regulator circuit 114 and IGBTs 140a,b, c, d of inverter circuit 118 in each separate dc--dc convertercircuit module 112-1 are switched on in a phased relationship withrespect to an adjacent module 112-2 . . . 112-n. For a plurality of nmodules, the phase relationship is equal to 360° /n.

In the case of four modular dc--dc converter sections as shown in FIG.1, the activation of each section or module 112-1 . . . 112-4 precedesthe activation of a successive module by 90°. A 4-phase oscillator 135generates the clock pulses separated by 90° to the modules. The phasedactivation of the separate modules 112-1, 112-2, 112-3 and 112-4 resultsin a frequency of the overall power supply output voltage ripple whichis four times higher than the frequency of the module output voltageripple. This permits a four times lower value of capacitance for theoutput filter capacitors 161 and 162 than would be necessary withoutphased activation.

A feedback control circuit shown generally at 130 in FIG. 1, controlsthe output voltage of the overall power supply in accordance with wellknown techniques. An analog voltage is generated by control circuit 130based on the difference between the desired output voltage, or powersupply voltage setpoint, and the actual output voltage as measured byvoltage divider 180. This analog voltage becomes the voltage setpoint toanother feedback control circuit in each dc--dc converter module. Aswill be explained more fully with reference to FIG. 2, this modulefeedback control circuit generates an error voltage which depends on thedifference between its setpoint and the derived output voltage of eachmodule. This error voltage is converted in the module control circuit toan enable signal having a variable duty cycle as a function of the errorvoltage. This enable signal is then applied by the control circuit tothe enable leads of the IGBT transistors 124a, b. The varying of the onand off times of IGBTs 124a, b maintains the output voltage from thedc--dc converter 112-1 at a controlled level under varying conditions ofload.

A ramp generation circuit 131 is part of the overall power supplyfeedback control circuit 130. The ramp generator slows down the voltagesetpoint applied to the feedback control summing means to avoidovershoots of the power supply output voltage. The ramp generator isneeded during step increases in power supply voltage setpoint or afterthe end of the cutback interval which occurs because of load arcing. Theramp time is on the order of 10 ms for an output power rating of 400 kW.

The operation of a single modular section, for example, 112-1 isdescribed with reference to FIG. 2. After rectification and filtering bythe three phase input rectifier and filter 100 shown in FIG. 1, a dcvoltage of 650 volts appears across input lines 201 and 202,respectively, as shown in FIG. 2. Lines 201 and 202 correspond to nodes101 and 102, respectively of FIG. 1. An input decoupling circuit 216comprises an inductor 222 in parallel with a resistor 220. The inputdecoupling circuit 216 provides a means for maintaining a substantiallysmooth current with low ripple from the output of the 650 volt sourceconnected across nodes 201 and 202. The input decoupling circuit 216also provides a means for preventing interactions between the modules112 which are connected in parallel across the 650 volt source. Theinput decoupling circuit 216 presents a relatively high impedancecompared to the impedance of the input capacitor 225 at the regulatorswitching frequency and above. As a result, it effectively forces thehigh frequency current pulses drawn by the regulator to flow fromcapacitor 225 and not from the input rectifier/filter 100 of FIG. 1.This substantially reduces electromagnetic interference caused by thepower supply since the currents in the cables between the inputrectifier/filter and the dc--dc converter modules are smooth with lowripple.

The input decoupling circuit 216 also damps out ringing. It will beappreciated that the input decoupling circuit 216 permits paralleloperation of the modules. 112-1 . . . 112-4 of FIG. 1 without largeinteractions. The input decoupling circuit 216 provides a means fordamping the resonance between cables and capacitors of differentmodules. The resistor 220 provides damping to the series resonantnetwork formed by the inductance of the cables between modules andcapacitance of the input capacitors in the different modules.

Inductor 222 is connected in parallel with the resistor 220 whichprovides a damping,means for suppressing oscillation in current in thetwo circuit input lines 201 and 202. The inductor 222 also provides ameans for limiting the rate of change of the current. Inductor 222 hasan input end and output end inserted in the circuit line from 201. Acapacitor 225 provides a charge storage means connected across the twocircuit lines 201 and 202 on the output side of the inductor 222 towardthe switching means 224a, 224b. This configuration insures that the linecurrent on the input end of the current rate of change limiting inductorincreases at a small rate from a first value to a high value when theswitching means 224a, 224b of the voltage regulator conduct current. Theline current on the input end of the current rate of change limitinginductor 222 decreases back to a first level at a small rate when theswitching means 224a, 224b of the regulator are nonconductive. Thecapacitor 225 absorbs all current flow through the current rate ofchange limiting inductor 222 when the switching means 224a, 224b arenonconductive. The capacitor 225 also adds to the current through thecurrent rate of change limiting inductor 222 to meet the currentrequirements of the voltage regulator switching means when the switchingmeans 224a, 224b are conducting.

Each dc--dc convertor module also includes a voltage regulator circuitmeans 214 for providing a pulsating voltage of controlled duty cyclefrom an unregulated dc input voltage on lines depending from nodes 201and 202 respectively. The voltage regulator circuit means 214 includes afree wheeling diode 226, voltage regulator switching means 224a, 224beach having a control lead, and emitter lead connected to a first outputterminal of the unregulated dc input voltage and having a collector leadconnected to the anode of free wheeling diode 226. The cathode of freewheeling diode 226 is connected through the input decoupling network 216to a second terminal of the unregulated dc input.

Inductor 228 acts as a means for smoothing current having a first endconnected to the common node of the voltage regulator switching means224a and 224b and to the anode of free wheeling diode 226. The inductormeans 228 provides a means for filtering the pulsating voltage createdby the switching means 224 and creates a smooth direct current forapplication to the inverter circuit means 218.

Referring to FIG. 2, the inverter circuit 218 provides a means forconverting the direct current from inductor 228 into a high frequencysquare wave current in the primary 252 of transformer 250. The invertercircuit 218 in a preferred embodiment comprises first and secondparallel circuit branches. The first circuit branch comprises IGBTs 240aand 240b, respectively, connected in series. A second branch comprisesIGBTs 240c and 240d, connected in series. Both circuit branches areconnected in parallel to form a bridge network. The IGBTs act as aswitching means, each having a collector lead, a control lead and anemitter lead. The IGBTs 240a and 240c each have their collectorsconnected to the positive inverter bus 241a which supplies a source ofdirect current. The emitter leads of IGBTs 240b and 240d are eachconnected to the negative inverter bus 241b which returns the directcurrent to inductor 228.

Also connected in parallel with the inverter circuit 218 betweenpositive bus 241a and negative bus 241b are the series connectedcapacitor 243 and diode 242 which, combined with resistor 244, form theinverter clamping means which is explained later. The primary winding252 of transformer 250 is connected across the output leads of theinverter. The four IGBTs are alternately activated in diagonal pairs(IGBTs 240a, 240d and IGBTs 240b, 240c) by voltages applied to theircontrol leads such that each output lead of the inverter is alternatelyconnected to positive and negative buses 241a, 241b of the invertercircuit 218. A short overlap time of approximately two microsecondsoccurs at each polarity transition where all four IGBTs are conducting.A substantially square wave alternating current is developed through theprimary 252 of the transformer 250.

Referring to FIG. 3, the inverter clamping network formed by a 2microfarad (μF) capacitor 343, diode 342, and 8 ohm resistor 344functions to limit the voltage between the positive and negative buseswhich feed the inverter, and thereby limit the voltage across the IGBTswitches, during change in polarity of the inverter output. FIG. 3 showsa portion of the inverter and regulator power circuit driving anequivalent representation of the step up transformer load. Inductor 355represents the transformer leakage inductance and voltage source 356represents the secondary voltage reflected to the primary. The magnitudeof voltage source 356 is substantially equal to the magnitude of themodule dc output voltage divided by the transformer turns ratio. Thepolarity of voltage source 356 is as shown in FIG. 3 when the directionof current i_(p) is as indicated in FIG. 3. It changes to the oppositepolarity when the current i_(p) changes to the opposite direction.

The principles of the inverter clamping network are best described byreferring to the waveforms during transition of polarity of the inverteroutput as shown in FIG. 4. For an inverter output frequency of 10 kHz,these waveforms occur every 100 microseconds during transition frompositive to negative inverter output polarity. The same waveforms but ofopposite polarity also occur every 100 microseconds during transitionfrom negative to positive inverter output. The transitions are spaced 50microseconds apart to create the substantially square current waveformsout of the inverter.

Referring to FIG. 4, prior to time t_(a) IGBT switches 340a and 340d areon and conducting and transistors switches 340b and 340c are off andnonconducting. The current I_(L) in inductor 328 flows through switches340a and 340d and through the equivalent load formed by the inductor 355and voltage source 356. At time t_(a), switches 340b and 340c are alsoturned on which results in zero voltage across the equivalent load andthe beginning of the decay of current i_(p) to zero which occurs at timet_(b). Between times t_(b) and t_(c), all four switches 340a, 340b,340c, and 340d are conducting current of magnitude I_(L) /2. At timet_(c), switches 340a and 340d are turned off and they becomenonconducting. The current I_(L) in inductor 328 is now forced to flowthrough the capacitor 343, which is precharged to 650 V through resistor344, and diode 342. This applies-650 V across the equivalent load whichresults in the buildup of current i_(p) in the negative direction. Thevoltage across capacitor 343 and the current i_(p) through theequivalent load continue to increase until the magnitude of i_(p)reaches the value I_(L) at time t_(d). At this time, t_(d), the diode342 becomes reverse biased and capacitor 343 begins discharging back to650 V through resistor 344 in preparation for the next polaritytransition one half cycle later.

It will be appreciated that an alternate current path is provided forcurrent flow as current builds up from zero in the inductive load 355. Acapacitor 343 is connected from one of the two input circuit lines andone lead of a voltage source capacitor 325 to a circuit node between aresistor 344 and diode 342, so that as current builds up from zero to amaximum value, excess current flow is from one of the two input circuitlines through the capacitor 343 and diode 342 to the other of the twoinput circuit lines. Voltage across the two circuit lines is clamped toapproximately the voltage of the voltage source 325. This inverterclamping network consisting of capacitor 343, diode 342, and resistor344 is superior to known prior art because the fast varying currentsoccur only within capacitor 343, diode 342, and the inverter IGBTswitches 340a, 340b, 340c, and 340d. Capacitor 343 and diode 342 arelocated very close to the inverter IGBTs which minimizes the strayinductance of the loop formed by the capacitor, diode, and IGBTs. As aresult, the voltage transient spikes appearing across the IGBTs, whichare created by this inductance and the fast change of currents, areminimized.

Referring again to FIG. 2, the inverter circuit also includes a meansfor simultaneously turning on all of the IGBT switch means 240a-240d forshorting and thereby isolating the primary 252 of transformer 250 whenan incipient gun arc is sensed. This minimizes the amount of powersupply current which passes through the load during load arcs.

It will be apparent to those skilled in the art that snubber networks,each consisting of a resistor, diode, and capacitor, may be needed inconjunction with each IGBT in the voltage regulator 214 and inverter218. These snubber networks are not shown in FIGS. 1, 2, and 3 forreasons of clarity. Designs of snubber networks for IGBTs are wellknown.

A one turn voltage sense winding 255 is wound on the same transformercore as primary 252 and secondary 254. Voltage sense winding 255provides a means for sensing an ac voltage proportional to the acvoltage across the secondary 254 of transformer 250. The ac voltageinduced in sense winding 255 is then rectified and filtered by an outputsimulator means 290. In addition, the output current of the module issensed by the dc current sensor 280 and used to control a current sourcein parallel with the filter capacitor 293 in the output simulator 290.The current source represents the current source characteristic of anion source, and specifically an electron beam gun. The capacitance valueof filter capacitor 293 and the value and range of current source 292are scaled so that the output of the output simulator 290 accuratelyrepresents the voltage level and circuit dynamics of the output of thedc--dc converter module and is consistent with the voltage levels usedin the feedback control circuit. Implementation of the output simulator290 is done with well known electronic circuit techniques. Use of theone turn sense winding 255 and the output simulator permit generation ofthe module output voltage feedback signal without requiring high voltageisolation circuitry which would be needed if the module output voltagewas measured directly.

The output simulator 290 comprises a diode rectifier means 291responsive to the ac voltage produced by the sense winding 255 toproduce a dc voltage on two output lines. This dc voltage isproportional to the dc output voltage of the power supply module 112.The output simulator means 290 also includes a filter capacitor 293connected across the two output lines from the rectifier means 291. Acontrolled current source means 292 is also connected across the twooutput lines for applying the output voltage of the current sensingmeans 280 to produce a current proportional to the output current of thepower supply module. The controlled current source means 292 drawscurrent out of the two output lines to create an accurate simulation ofpower supply module output voltage.

The output of the output simulator 290 is then applied to a voltagesumming means 266. In the summing means, voltage from the outputsimulator 290 is subtracted from a set point voltage. The set pointvoltage is a reference voltage for controlling the amplitude of thedc--dc converter output voltage. The set point voltage enables theamplitude of the converter output voltage to be controlled to a desiredlevel. The output of the voltage summing means 266 is a differencevoltage, that is, the difference between the set point voltage and thevoltage output simulator.

The difference voltage is amplified and filtered in accordance withknown techniques in the compensator 268. The amplified voltage from thecompensator 268 is then compared to a clock generated sawtooth voltagein a comparator 270. The sawtooth voltage is generated by a sawtoothgenerator means 272 and is synchronized to the clock pulse of thesystem. The output of the comparator is a logic level enable signal (+5Vfor example) which has an enable and a disable state. The output of thecomparator 270 and the clock logic signal feed the regulator logic anddriver circuit 274 which distributes the enable signal to the regulatorIGBTs 224a and 224b. Logic and driver circuit 274 also provides themeans for alternately enabling IGBTs 224a and 224b. This logic anddriver circuit 274 is implemented using well known electronictechniques. The enable signal has a variable ratio of the time duringwhich the voltage is high to the time during which the voltage is low asa function of the error voltage output of compensator 268. The enablesignal is then applied to the enable leads of the IGBT switch means 224aand 224b of the voltage regulator circuit 214. This provides dynamicregulation of the dc voltage on the output of the power supply inaccordance with varying loads being sensed by the output simulator 290and sense winding 255.

In accordance with another aspect of the present power supply, a dccurrent sensor means 280 is provided on the negative output lead of thepower supply. There, the sensed output current is compared with apredetermined threshold indicative of current conditions during anincipient gun arc. The sensed current and current threshold are comparedin a voltage comparator 282. The voltage comparator produces an outputsignal when current above a predetermined threshold is sensed and anincipient gun arc is present. This in turn sends a signal to a cutbacktimer 284 which generates a pulse ranging from 50 ms to 200 ms and whichin turn activates the inverter logic and driver circuit 286. When a gunarc is sensed, the inverter logic and driver circuit 286 simultaneouslyactivates all four IGBTs 240a-240d which shorts out the primary 252 andthereby terminates current let through from the primary side to the loadside. The cutback timer 284 also simultaneously sends a signal to theregulator logic and driver circuit 274 which turns off the regulatorswitch means 224a and 224b.

While the invention has been described in connection with what ispresently considered to be the most practical and preferred embodiments,it is to be understood that the invention is not limited to thedisclosed embodiment, but on the contrary, is intended to cover variousmodifications and equivalent arrangements included within the spirit andscope of the appended claims.

What is claimed is:
 1. Circuit means for generating a ground levelvoltage feedback signal for controlling the output voltage of one of aplurality of dc--dc converter modules having their outputs connected inseries to form a supply output lead, each module including a switchingmeans for producing a pulsating voltage of controlled duty cycle, aninductor means for converting said pulsating voltage into a smoothdirect current, an inverter means for producing from said direct currentan alternating current through the primary of a transformer, saidtransformer having at least one secondary winding inductively coupled tosaid primary winding for producing an output voltage of said module,said circuit means for generating the output voltage feedback signalcomprising:a separate transformer winding inductively coupled with saidprimary and said secondary for producing an ac voltage whose magnitudeis proportional to the dc output voltage of said dc--dc convertermodule; rectifier means responsive to said ac voltage produced by saidseparate transformer winding having a connection with a capacitor forproducing a dc voltage on two output lines which voltage is proportionalto the dc output voltage of said dc--dc converter module; currentsensing means connected to said supply output lead for producing anoutput voltage proportional to the current flowing through the output ofall of said series connected dc--dc converter modules; controlledcurrent source means connected across said two output lines for applyingthe output voltage of said current sensing means to produce a currentproportional to the output current of said dc--dc converter modulewherein said controlled current source means draws current out of saidtwo output lines to create an accurate simulation of the module outputload.
 2. A circuit means for decoupling a power supply output from loadgenerated arcs comprising:a power supply including a plurality of dc--dcconverter modules each having two input lines and two output lines, saidmodules linked in parallel on their input lines and in series on theiroutput lines; regulating means, provided in each module, for producing acontrolled source of direct current; inverter means for generating asquare wave current of alternating polarity from said controlled sourceof direct current; transformer means for stepping up an applied voltageand for stepping down said square wave current from said inverter means;output rectifier and filter capacitor means for rectifying and filteringsaid square wave current from said transformer; inductor means disposedin a power supply output line for limiting current rate of changebetween said plurality of said series connected output lines of saiddc--dc converter modules and a supply output line; resistor dampingmeans connected in parallel with said inductor means, said resistordamping means for reducing oscillation caused by resonance between saidinductor and said filter capacitor.
 3. An apparatus according to claim 2wherein said inductor means for limiting current rate of change and saidresistor damping means for reducing oscillation, connected in parallelwith said inductor means, are inserted in each output line of saidplurality of dc--dc converter modules.
 4. An apparatus according toclaim 3 wherein said plurality of dc--dc converter modules have theiroutput lines connected in parallel.
 5. A power supply for producing acontrolled high voltage output when operating with rapidly changingloads such as when load arcing is present comprising;a means forrectifying a source of multiphase ac voltage and for producing anunregulated dc voltage on positive and negative out leads; a pluralityof dc--dc converter modules connected in parallel across said positiveand negative dc output leads of said means for rectifying, each of saidconverter modules including a regulator means for producing a pulsatingvoltage of controlled duty cycle, an inductor means for smoothing saidpulsating voltage into a smooth direct current, inverter switch meansfor converting said direct current into high frequency square wavecurrent of alternating polarity, transformer means having a primarywinding for receiving said high frequency square wave current and forstepping up the applied voltage across said primary winding to a highvoltage level across a secondary winding, output rectifier and filtercapacitor means for converting the high frequency, high voltagedeveloped across said secondary into a high direct voltage on a positiveand negative output lead of a first output rectifier and filter of afirst dc--dc converter module connects to the negative output lead of asecond output rectifier and filter of a second circuit module such thatthe dc output voltage produced by all circuit modules are added and thesum of said voltage is developed across a positive and negative outputline of the power supply; output simulator means including a voltagesense winding means inductively coupled with each primary and secondarywinding of said transformers in said plurality of dc--dc convertermodules and for producing a simulated output voltage proportional to thedc--dc converter module output voltage; and control means for activatingeach successive dc--dc converter module in a phased relationship, 360°/n out of phase with an adjacent dc--dc converter module, where n is thenumber of dc--dc converter modules in said plurality of dc--dc convertermodules, and for sensing in each of said dc--dc converter modules saidsimulated voltage and including means for regulating said dc voltage asa function of the difference between said simulated voltage and adesired output voltage.
 6. A power supply according to claim 5 whereinsaid plurality of dc--dc converter modules each have their respectiveoutput leads connected in parallel.
 7. A power supply according to claim5 wherein each of said plurality of dc--dc converter modules furthercomprises:a regulator switching means having a collector lead, a controllead and an emitter wherein said emitter is connected to said negativedc output of said rectifier, said collector is connected to the anode ofa free wheeling diode and to a first lead of an inductor; an invertermeans having first and second input leads wherein said first input leadconnects to a second input lead of said inductor and said second inputlead connects to said free wheeling diode cathode and to the positive dcoutput lead of said rectifier; control means for sequentially enablingsuccessive regulator switch means of each successive converter moduleequally out of phase with an adjacent converter module.